Improving the technical characteristics of radio receivers. Encyclopedia of radio electronics and electrical engineering

Encyclopedia of radio electronics and electrical engineering / radio reception

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Using VHF quartz filters, Schottky diodes and high-power RF transistors, it is possible to significantly improve receiver parameters such as linearity and image selectivity. Eight ways to improve radio receivers are described, including choosing a high intermediate frequency, using separate AGC and gain, using push-pull frequency converters, using stages for double balanced frequency converters with Schottky diodes, and optimal distribution of AGC across receiver stages.

Despite the fact that radios began to be developed at the dawn of electronic technology, there are still ways to further improve them. New components such as meter-wavelength crystal filters, pin diodes, and high-power high-frequency transistors make it possible to break away from some established concepts and develop receivers with less distortion, better image selectivity, and high linearity.

Particularly tangible benefits can be realized in the range of 2-30 MHz, however, many of the proposed methods are applicable to receivers operating at other frequencies.

The first step in designing a receiver is to draw up a block diagram, on which the expected noise figures and losses are noted for each block (losses are also sources of additional noise). This makes it possible to calculate the noise figure of the entire receiver. For example, in the block diagram of the receiver shown in Fig. 1, the noise figure, determined by summing noise and loss, is 8 dB.

Rice. 1. The block diagram of the receiver allows you to determine the gains of the individual stages and the noise figure of the receiver (click to enlarge)

The noise figure of the entire receiver is determined by summing the noise figure, gain, and loss (in decibels) of the individual stages. To obtain a wide dynamic range, the gain must be as low as necessary to compensate for the loss.

Each stage needs to be optimized in terms of dynamic range and noise figure. The maximum dynamic range is obtained if the gains of the RF and IF stages have the minimum value necessary to compensate for losses. As can be seen from the block diagram, losses of 0,5 dB in the input circuit and AGC attenuator, 6,5 dB in the frequency converter and 4,5 dB in the IF filter are compensated by a gain of about 11 dB in the RF amplifier. It should be noted that the second frequency converter is the most sensitive to overloads, since the minimum bandwidth of the crystal filter is ±3,5 kHz, and, therefore, in this stage, higher voltages are concentrated in a narrow frequency band.

After selecting the main parameters of the block diagram, the developer can proceed to the design of individual cascades. It is at this stage that the benefits of the new components can be realized. Consider the sequence of ways to improve the receiver.

1. To obtain better selectivity over the image channel, the intermediate frequency must be higher than the received frequency range

In the past, in double or triple conversion receivers, each of the two or three intermediate frequencies, respectively, was below the frequencies of the received band, and receiver selectivity was determined mainly by circuits operating at the lowest intermediate frequency (often 455 kHz). This is explained by the fact that the components available at that time could provide the required selectivity only at low intermediate frequencies. However, at a low first intermediate frequency, the problem of attenuating the image channel noise becomes more difficult. The noise frequencies acting at the input, after the converter, to which the local oscillator voltage is applied, can fall into the passband of the IF. In the case of an IF of 1 MHz, the attenuation of the image channel interference, although it is 80 dB at the lowest receive frequency (2 MHz), drops to 30 dB at 30 MHz. For example, in the case of receiving a signal with a frequency of 30 MHz, the interference on the image channel has a frequency of 32 MHz, which is close to the frequency of the received signal and cannot be sufficiently attenuated by the input filter. At the same time, when receiving at a frequency of 2 MHz, the interference frequency of 4 MHz is twice as high as the input frequency, which provides good selectivity over the image channel. To attenuate the interference on the image channel, which have frequencies close to those received, the developers tried to use tracking bandpass filters in the preselectors, which increased the cost of the receiver.

The local oscillator must be tuned in a range equal in width to the frequency range of the input signals. So, in a receiver with a range of 2-30 MHz, the local oscillator coverage ratio should be 1:15. This overlap ratio may require complex mechanical arrangements to accurately match the input and local oscillator circuit settings. Using currently available quartz filters in the range of meter waves (30 - 120 MHz) in the IF cascades, the above problems can be solved. By selecting an intermediate frequency above the frequencies of the operating range, it is possible to use an elliptical low-pass filter with a cutoff frequency of, for example, 2 MHz in a receiver with a range of 30-31 MHz. In this case, interference with frequencies above the operating range is attenuated by 80 dB, and the selectivity over the image channel does not depend on the frequency of the received signals. The same filter will provide attenuation of the local oscillator radiation, which allows you to place several receivers at a close distance from each other. When the intermediate frequency is, for example, 40 MHz, the local oscillator should cover the range 42-70 MHz (in a receiver with a range of 2-30 MHz); therefore, the overlap ratio is less than 1:2. This greatly simplifies the design of the local oscillator and reduces the likelihood that the interaction of harmonics of the local oscillator with the input signals in the frequency converter will lead to the formation of interference falling into the receiver bandwidth.

2. The use of separate stages for AGC and for amplification in order to reduce distortion.

In the past, vacuum tubes were used for both amplification and AGC. However, due to the non-linearity of the lamp characteristics, intermodulation distortion occurred when the AGC voltage was applied. The same is true when using bipolar and field-effect transistors. If the amplification and AGC are carried out in separate stages, then it is possible to provide the optimal mode for each of them. So, for example, for AGC, you can use an attenuator on pin diodes. connected between the input low-pass filter and the RF amplifier, as shown in Fig.1. The diode attenuator must have constant input and output impedances, otherwise any change in the load impedance will change the characteristics of the filter, and a change in the source impedance driving the amplifier will change the noise and distortion in it. On fig. 2 shows the attenuator, which is a conventional double T-bridge on pin diodes. The input and output impedances of such an attenuator are kept constant. For this purpose, a differential amplifier is used, which provides an appropriate redistribution of currents in the outputs of the attenuator (the sum of the collector currents must be unchanged).

Rice. 2. Five pin diodes connected in a double T-bridge circuit act as an attenuator. To keep the input and output impedances constant, the sum of the collector currents of the transistors must be kept constant.

3. The use of push-pull RF amplifiers on powerful transistors with deep feedback to reduce distortion

In most older receivers, only a few tubes were considered sufficiently linear for use in Class A mode input amplifiers. The designers used the properties of these tubes to achieve low intermodulation distortion. High-power linear high-frequency transistors are currently being produced, which, operating in high DC modes with strong current and voltage feedback (which is rarely used in practice), can provide even better linearity than lamps. On fig. 3 shows a diagram of such an amplifier, assembled on powerful linear transistors of the decimeter wave range.

Rice. 3 (click to enlarge). On two powerful high-frequency transistors, you can assemble an amplifier with good linearity. To linearize the amplifier, feedback is used, created by an emitter resistor without a shunt capacitor, collector-base resistors, and collector-base transformers. The curves illustrate the decrease in non-linearity.

A push-pull amplifier attenuates second-order non-linearity products by 40 dB relative to a single-ended one. The gain depends on the depth of feedback and in the variant of Fig. 3 equals 11 dB. The introduction of feedback reduces the gain by 40 dB while expanding the dynamic range. The amplifier uses three types of feedback: current feedback is provided by a 6,8 ohm emitter resistor without a bypass capacitor; a 330 ohm resistor connected between collector and base without shunt capacitor provides voltage feedback. Since these feedbacks change the input and output impedances, a transformer feedback is also introduced, due to which the output and input impedances are equal to 50 ohms. At the same time, the a.s.v.s. amplifier does not exceed 1,2 in the frequency range from 100 kHz to almost 200 MHz. The advantages of this new type of RF amplifier are best illustrated by its characteristic shown in Fig. 3. With an input power of -27 dBm (two sinusoidal signals with amplitudes of 20 mV each), the gain is 12 dB. With such an input signal, the level of second-order intermodulation products (f1±f2) in a single-cycle cascade does not exceed -65 dB, and third-order products (f1±2f2) -100 dB. In the push-pull amplifier, the second-order non-linear products are further reduced to -105 dB. The third order non-linearity product level reaches the desired output level at +22 dBm input power.

4. Application of double balanced frequency converters with Schottky diodes

The advantages of push-pull converters over single-cycle converters are known (high sensitivity, low distortion), but the high cost prevents their wide distribution. Currently low-noise conversion diodes on hot carriers (Schottky diodes) are produced at an affordable price. It should be noted that double balanced FET converters are also currently being produced. Such converters provide good suppression of third-order non-linearity products, but due to poor matching of field-effect transistors, the attenuation of second-order non-linearity products in them is 20-30 dB worse than on Schottky diodes. In addition, FETs limit signals at lower levels than Schottky diodes.

The main advantage of Schottky diode mixers is that they allow better matching compared to conventional silicon or germanium diodes. Such mixers can operate at higher voltage from the local oscillator. Schottky diode noise lacks the 1/f2 component that prevents silicon diodes from being used at low frequencies. In order to optimize the characteristics of the frequency converter, the circuits shown in fig. 4, a and b. Sometimes the converter contains up to 64 diodes (16 in each section). The second converter in application according to the block diagram of fig. 1 handles larger signals than the first, so it should have a wider dynamic range. In the converter according to the scheme of fig. 4, and this is achieved by including series resistors and using a push-pull circuit.

Rice. 4. By including resistors in series with the diode bridges in a push-pull frequency converter, you can maintain the dynamic range at high signal levels (a). The transformer in the lower circuit serves to suppress parasitic signals.

It should be noted that series resistors increase mixer losses from 6,5 to 8 dB. In the converter according to the scheme of fig. 4b, a hybrid transformer is used to suppress side channel interference.

5. Use of quartz filters with low losses to obtain high selectivity in the cascades of the first intermediate frequency (meter waves) and effective attenuation of interference in the image channel.

Until recently, it was impossible to mass-produce quartz filters with high selectivity and low insertion loss. On fig. 5a shows the frequency response typical of modern quartz filters. Since the attenuation of the image channel interference between the first and second intermediate frequencies is determined by the slope of the filter's frequency response, the image channel selectivity can be as high as 80 dB. The price of one such filter was recently $400, and now in serial production it has dropped to $50. Old-style mechanical filters (with a magnetostrictive converter) introduced strong intermodulation distortion due to the nonlinearity of the converter. In modern mechanical filters, piezoelectric transducers are used to reduce non-linearity. Similar effects can occur in quartz filters if the ferromagnetic core of the input transformer saturates at low signal levels. To reduce the nonlinearity, you can apply the scheme of Fig. 5 B. The tests are carried out with two signals with an amplitude of 1 V applied to the 50-ohm filter input; while the level of the spurious signal should not exceed -80 dB.

Rice. 5 (click to enlarge). Using crystal filters, you can achieve low losses and steep edges in the frequency response (the lower curve is stretched for clarity relative to the middle frequency). The circuit shown in the figure uses the same type of filters tuned to the same operating frequency.

6. Double frequency conversion, together with non-tunable low-pass filters, allows you to adjust the bandwidth without changing the steepness of the slope of the frequency response.

Obtaining a rectangular frequency response of the IF with the use of narrow bandpass filters has always been a serious problem. The new double-inverted input spectrum scheme can apply low-pass filters, while the slope of the frequency response of the IF is independent of the bandwidth. An additional advantage of low-pass filters is that the settling time is half that of band-pass filters. This eliminates unwanted fluctuations in the filters in the case of pulsed signals. The essence of the method is illustrated by the diagram (Fig. 6).

Rice. 6 (click to enlarge). The bandwidth of the IF receiver depends on the frequency offset between the two local oscillators operating at the second intermediate frequency. The input signal is frequency-converted twice in the 52-64 kHz range, and steep filters at 64 kHz form the fronts of the frequency response of the IF.

The selectivity of the receiver is determined mainly by the path of the second intermediate frequency 525 kHz. The bandwidth of the second intermediate frequency, and therefore the bandwidth of the receiver as a whole, can be set within 150 Hz-12 kHz. In this case, the choice of the bandwidth is not carried out by replacing the filter, but by adjusting the frequency shift between the two local oscillators. A 525 kHz signal with a maximum bandwidth of, say, ±6 kHz (510-531 kHz) enters the frequency converter initially at 467 kHz LO, resulting in a signal spanning 52 (525-6-467) to 64 kHz (525+6-467). The resulting signal is fed into a low pass quartz filter whose frequency response has a sharp rolloff at 64 kHz (this rolloff forms one of the fronts of the IF frequency response). The specified filter with a fixed cutoff frequency is adjusted only once. Then the signal spectrum with a bandwidth of 52-64 kHz is again transferred to the center frequency of 525 kHz and again fed to the converter with a local oscillator frequency of 583 kHz. In this case, the signal returns to the range of 52-64 kHz, but with an inverted spectrum (spectrum components that were previously at the 64 kHz bandwidth boundary are now 12 kHz below this boundary). A filter with a cutoff frequency of 64 kHz suppresses signal components that were at the 52 kHz boundary during the first conversion. The signal obtained in this way, filtered with high selectivity, is again transferred over the spectrum to a frequency of 525 kHz and detected.

It should be noted that the edges of the frequency response of the IF are kept unchanged, and the bandwidth is reduced by adjusting the frequency shift between the two local oscillators. So, for example, with a bandwidth of 2 kHz, the local oscillators are tuned to frequencies of 462 kHz (525 + 1-64) and 588 (525-1 + 64). Due to the fact that the band edges are formed by the low-pass filter, the frequency response is close to rectangular even at a bandwidth of 150 Hz. The described method ensures the symmetry of the phase response or group delay characteristics with respect to the center frequency. Crystal or mechanical filters commonly used in IF are Chebyshev filters with a non-linear phase response. At the same time, low-pass filters of the Bessel type can provide the required linearity.

7. Among the factors that degrade the dynamic range of the receiver, it is necessary to take into account the noise sidebands of the local oscillator

The noise sidebands of the LO spectrum can significantly degrade the dynamic range of the receiver due to an effect called blocking. LO noise can interfere with strong input signals close in frequency to the received signal, resulting in noise in the IF passband that interferes with the wanted signal, reducing the signal-to-noise ratio. Strong blocking distortion can occur at signal levels well below the 3dB compression threshold (another dynamic range parameter). The 3 dB compression threshold corresponds to the appearance of noticeable cross modulation and usually occurs at higher signal amplitudes than the blocking effect. From fig. In Figure 7, as an example, it can be seen that with a sideband noise spectral density of 145 dB/Hz (20 kHz offset from the LO center frequency) and a receiver noise figure of 10 dB, a receiver blocking of 3 dB occurs at an input voltage of about 50 mV, while how the 3 dB compression threshold corresponds to a signal amplitude of about 1 V.

Rice. 7. Input voltage corresponding to a 3 dB blocking effect as a function of the local oscillator sideband noise that is converted by the input signals and the total receiver noise figure.

When using a frequency synthesizer as a local oscillator, it is also necessary to eliminate spurious signals, since they, like noise sidebands, can degrade the performance of the receiver.

8. Proper distribution of AGC across receiver stages to obtain maximum dynamic range

The dynamic range of the receiver depends on the lowest signal level at which the AGC voltage is applied to the RF attenuator. Until the signal level in the antenna reaches a value corresponding to the signal-to-noise ratio of 48 dB, the AGC should operate only in the IF (Fig. 8).

Rice. 8. AGC circuits increase the signal-to-noise ratio (vertical difference between two curves). To obtain a wide dynamic range, the AGC input attenuator should only be activated when the signal-to-noise ratio reaches 48 dB.

After that, the AGC attenuator should come into action, which protects the second converter from overload. If the AGC attenuator starts to work at smaller signals, then not only will the signal-to-noise ratio decrease, but the stability of the AGC may deteriorate. The AGC circuit must be carefully analyzed as a closed-loop system, for example, using a Nyquist hodograph, in order to optimize its parameters.

Literature

1. Eight ways to better radio receiver design, pp. 87-91

Publication: N. Bolshakov, rf.atnn.ru

See other articles Section radio reception.

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