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ENCYCLOPEDIA OF RADIO ELECTRONICS AND ELECTRICAL ENGINEERING Input circuits and RF receiver. Encyclopedia of radio electronics and electrical engineering
Encyclopedia of radio electronics and electrical engineering / radio reception As we already found out in the first chapter, in order to increase the sensitivity and real selectivity of a heterodyne receiver, the input circuit must provide a power transfer coefficient close to unity in the operating frequency range and as much as possible attenuation of out-of-band signals. All this is the properties of an ideal band-pass filter, therefore, the input circuit must be implemented in the form of a filter. The often used single-loop input circuit is the worst to meet the requirements. To increase selectivity, it is necessary to increase the loaded quality factor of the circuit, weakening its connection with the antenna and mixer or URF. But then almost all the power of the received signal will be spent in the circuit and only a small part of it will pass into the mixer or URF. The power transfer coefficient will be low. If, however, the circuit is strongly connected to the antenna and the mixer, the loaded quality factor of the circuit will drop and it will slightly attenuate the signals of stations adjacent in frequency. But next to the amateur bands, very powerful broadcasting stations also work. A single input circuit as a preselector can be used on low-frequency KB bands, where signal levels are quite high, in the simplest heterodyne receivers. Communication with the antenna should be made adjustable, and the circuit itself tunable, as shown in Fig. 1. In the case of interference from powerful stations, you can weaken the connection with the antenna by reducing the capacitance of the capacitor C1, thereby increasing the selectivity of the circuit and at the same time increasing the losses in it, which is equivalent to turning on the attenuator. The total capacitance of capacitors C2 and C3 is chosen around 300 ... 700 pF, these coils depend on the range.
Significantly better results are obtained by band-pass filters matched at the input and output. In recent years, there has been a trend to apply switchable bandpass filters even at the input of wide-range professional communication receivers. Use octave (rarely), half-octave and quarter-octave filters. The ratio of the upper frequency of their bandwidth to the lower one is equal to 2, respectively; 1,41 (square root of 2) and 1,19 (fourth root of 2). Of course, the narrower the input filters, the higher the noise immunity of the wide-range receiver, but the number of switched filters increases significantly. For receivers designed only for amateur bands, the number of input filters is equal to the number of bands, and their bandwidth is chosen equal to the band width, usually with a margin of 10 ... 30%. In transceivers, it is advisable to install band-pass filters between the antenna and the antenna receive / transmit switch. If the transceiver's power amplifier is wide enough, as is the case with a transistor amplifier, its output may contain many harmonics and other out-of-band signals. A bandpass filter will help suppress them. The requirement for a filter power transfer coefficient close to unity is especially important in this case. The filter elements must be capable of withstanding reactive power several times the rated power of the transceiver's transmitter. It is advisable to choose the characteristic impedance of all band filters to be the same and equal to the wave impedance of the feeder 50 or 75 Ohm.
The classical scheme of the L-shaped band-pass filter is given in Fig. 2a. Its calculation is extremely simple. First, the equivalent quality factor Q = fo/2Df is determined, where fo is the middle frequency of the range, 2Df is the filter bandwidth. The inductance and capacitance of the filter are found by the formulas:
where R is the characteristic impedance of the filter. At the input and output, the filter must be loaded with resistances equal to the characteristic, they can be the input impedance of the receiver (or the output of the transmitter) and the antenna impedance. Mismatch up to 10...20% practically has little effect on the characteristics of the filter, but the difference between the load resistance and the characteristic resistance by several times sharply distorts the selectivity curve, mainly in the passband. If the load resistance is less than the characteristic one, it can be connected by an autotransformer to the tap of the L2 coil. The resistance will decrease in k2 times, where k is the turn-on ratio, equal to the ratio of the number of turns from the outlet to the common wire to the total number of turns of the coil L2. The selectivity of one L-shaped link may be insufficient, then two links are connected in series. Links can be connected either in parallel branches to each other, or in series. In the first case, a T-shaped filter is obtained, in the second, a U-shaped one. The L and C elements of the connected branches are merged. As an example, Fig. 2b shows a U-shaped bandpass filter. The elements L2C2 remained the same, and the elements of the longitudinal branches were combined into an inductance 2L and a capacitance C1 / 2. It is easy to see that the tuning frequency of the resulting series circuit (as well as the rest of the filter circuits) remained the same and equal to the middle frequency of the range. Often, when calculating narrow-band filters, the value of the capacitance of the longitudinal branch C1 / 2 turns out to be too small, and the inductance is too large. In this case, the longitudinal branch can be connected to the taps of the coils L2, increasing the capacitance by 1/k2 times, and the inductance is reduced by the same amount.
In RF filters, it can be convenient to use only parallel oscillatory circuits connected by one output to a common wire. The scheme of a two-loop filter with external capacitive coupling is shown in Fig.3. The inductance and capacitance of the parallel circuits are calculated by formulas (1) for L2 and C2, and the capacitance of the coupling capacitor should be C3=C2/Q. The switching coefficients of the filter outputs depend on the required input resistance Rin and the characteristic impedance of the filter R: k2=Rin/R. The turn-on coefficients on both sides of the filter can be different, providing matching with the antenna and the receiver input or transmitter output. To increase the selectivity, three or more identical circuits can be switched on according to the scheme of Fig. 3, reducing the capacitances of the coupling capacitors C3 by 1,4 times.
The theoretical selectivity curve of a three-loop filter is shown in Fig.4. The relative detuning x=2DfQ/fo is plotted horizontally, while the attenuation introduced by the filter is plotted vertically. In the transparency band (x<1), the attenuation is zero, and the power transfer coefficient is one. This is understandable if we take into account that the theoretical curve is constructed for lossless elements with an infinite design quality factor. A real filter also introduces some attenuation in the passband, which is associated with losses in the filter elements, mainly in the coils. The losses in the filter decrease with the increase in the constructive quality factor of the coils Q0. For example, at Q0 = 20Q, losses even in a three-loop filter do not exceed 1 dB. The attenuation outside the passband is directly related to the number of filter loops. For a two-loop filter, the attenuation is 2/3 indicated in Fig. 4, and for a single-loop input circuit it is 1/3. For the U-shaped filter Fig. 3b, the selectivity curve Fig. 4 is suitable without any correction.
A practical scheme of a three-loop filter with a bandwidth of 7,0...7,5 MHz and its experimentally measured characteristic are shown in Figs. 5 and 6, respectively. The filter is calculated according to the described method for the resistance R=1,3 kOhm, but was loaded on the input resistance of the heterodyne receiver mixer 2 kOhm. Selectivity increased slightly, but peaks and dips appeared in the passband. The filter coils are wound turn to turn on frames with a diameter of 10 mm with PEL 0,8 wire and contain 10 turns each. The withdrawal of the coil L1 to match the resistance of the antenna feeder 75 ohms is made from the second turn. All three coils are enclosed in separate screens (aluminum cylindrical "cups" from nine-pin lamp panels). Filter tuning is simple and comes down to tuning the circuits to resonance with coil trimmers.
Particular attention should be paid to the issues of obtaining the maximum constructive quality factor of the filter coils. One should not strive for special miniaturization, since the quality factor increases with the increase in the geometric dimensions of the coil. For the same reason, it is undesirable to use too thin a wire. Silvering the wire gives a noticeable effect only on high-frequency HF bands and on VHF with a constructive quality factor of the coil of more than 100. It is advisable to use litz wire only for winding coils in the ranges of 160 and 80 m. Lower losses in the silver-plated wire and litz wire are due to the fact that high-frequency currents do not penetrate into thickness of the metal, but flow only in a thin surface layer of the wire (the so-called skin effect). A perfectly conductive screen does not reduce the quality factor of the coil and also eliminates energy losses in the objects surrounding the coil. Real screens introduce some losses, so it is advisable to choose a screen diameter equal to at least 2-3 coil diameters. At the same time, the inductance also decreases to a lesser extent. The main purpose of screens is to eliminate parasitic connections between elements. It makes no sense, for example, to talk about obtaining an attenuation of more than 20 ... 30 dB if the filter details are not shielded and the signal can be induced from the input circuits to the output ones. The screen should be made of a well-conductive material (copper, aluminum is somewhat worse). Painting or tinning of the inner surfaces of the screen is not allowed. These measures provide exceptionally high quality factor coils, implemented, for example, in helical resonators. In the 144 MHz range, it can reach 700 ... 1000. Figure 7 shows the design of a 144 MHz dual-cavity bandpass filter designed for inclusion in a 75 ohm feedline. The resonators are mounted in rectangular screens 25X25X50 mm in size, soldered from sheet copper, brass or plates of double-sided foil fiberglass. The inner baffle has a connection hole measuring 6X12,5mm. Air tuning capacitors are fixed on one of the end walls, the rotors of which are connected to the screen. The resonator coils are frameless. They are made of silver-plated wire with a diameter of 1,5 ... 2 mm and have 6 turns with a diameter of 15 mm, evenly stretched to a length of about 35 mm. One output of the coil is soldered to the stator of the trimmer capacitor, the other to the screen. The taps to the input and output of the filter are made from 0,5 turns of each coil. The bandwidth of the tuned filter is slightly more than 2 MHz, the insertion loss is calculated in tenths of a decibel. The filter bandwidth can be adjusted by changing the size of the coupling hole and selecting the position of the coil taps.
On higher-frequency VHF bands, it is advisable to replace the coil with a straight piece of wire or tube, then the spiral resonator turns into a coaxial quarter-wave resonator loaded with a capacitance. The length of the resonator can be chosen about l / 8, and the length missing up to a quarter of the wavelength is compensated by a tuning capacitance. In particularly difficult reception conditions on the KB bands, the input circuit or filter of the heterodyne receiver is made narrow-band, tunable. To obtain a high loaded quality factor and a narrow band, the connection with the antenna and between the circuits is chosen to be minimal, and to compensate for the increased losses, a field-effect transistor amplifier is used. Its gate circuit shunts the circuit little and reduces its quality factor almost nothing. It is impractical to install bipolar transistors in URF due to their low input resistance and much greater non-linearity. The URCH scheme is shown in Fig. 8. A two-circuit tunable band-pass filter at its input provides all the required selectivity, therefore, a non-tunable low Q circuit L3C9, shunted by resistor R3, is included in the drain circuit of the transistor. This resistor selects the gain of the cascade. Due to the low amplification of the neutralization of the pass capacitance of the transistor is not required.
The drain circuit can also be used to obtain additional selectivity if the shunt resistor is omitted, and the drain of the transistor is connected to the tap of the loop coil to reduce the gain. The scheme of such an URCh for a range of 10 m is shown in Fig. 9. It provides a receiver sensitivity better than 0,25 μV. In the amplifier, double-gate transistors KP306, KP350 and KP326 can be used, which have a small throughput capacitance, which contributes to the stability of the URF with a resonant load.
The transistor mode is set by selecting resistors R1 and R3 so that the current consumed from the power source is 4 ... 7 mA. The gain is selected by moving the L3 coil tap and when the coil is fully turned on it reaches 20 dB. The L2 and L3 loop coils are wound on K10X6X4 rings made of 30VCh ferrite and have 16 turns of PELSHO 0,25 wire. Coils of communication with the antenna and the mixer contain 3-5 turns of the same wire. It is easy to introduce an AGC signal into the amplifier by applying it to the second gate of the transistor. When the potential of the second gate is reduced to zero, the gain decreases by 40...50 dB. Literature
Author: V.T.Polyakov; Publication: N. Bolshakov, rf.atnn.ru
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