We measure SWR: theory and practice. Encyclopedia of radio electronics and electrical engineering

Encyclopedia of radio electronics and electrical engineering / Antennas. Measurements, adjustment, coordination

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A device for measuring the quality of matching the feeder with the antenna (SWR meter) is an indispensable part of an amateur radio station. How reliable information about the state of the antenna economy does such a device provide? Practice shows that not all factory-made SWR meters provide high measurement accuracy. This is even more true when it comes to homemade designs. In the article brought to the attention of readers, an SWR meter with a current transformer is considered. Devices of this type are widely used by both professionals and radio amateurs. The article gives the theory of its work and analyzes the factors affecting the accuracy of measurements. It ends with a description of two simple practical designs of SWR meters, the characteristics of which will satisfy the most demanding radio amateur.

Some theory

If a homogeneous connecting line (feeder) with wave impedance Zо connected to the transmitter is loaded with resistance Zн≠Zо, then both an incident and a reflected wave appear in it. The reflection coefficient r (reflection) is generally defined as the ratio of the amplitude of the wave reflected from the load to the amplitude of the incident wave. The current reflection coefficients r and voltage ru are equal to the ratio of the corresponding quantities in the reflected and incident waves. The phase of the reflected current (with respect to the incident current) depends on the ratio between Zн and Zо. If Zн>Zо, then the reflected current will be antiphase to the incident one, and if Zн

The value of the reflection coefficient r is determined by the formula

where Rn and Xn are the active and reactive components of the load resistance, respectively. With a purely active load Xn = 0, the formula is simplified to r=(Rn-Zo)/(Rn+Zo). For example, if a cable with a characteristic impedance of 50 ohms is loaded with a 75 ohm resistor, then the reflection coefficient will be r = (75-50)/(75+50) = 0,2.

On fig. 1a shows the distribution of voltage Ul and current Il along the line for this particular case (losses in the line are not taken into account). The scale along the y-axis for the current is taken to be Z times larger - in this case, both graphs will have the same vertical size. The dotted line is the graphs of voltage Ulo and current Ilo in the case when Rн=Zо. For example, a line section of length λ is taken. With its longer length, the pattern will be cyclically repeated every 0,5λ. At those points of the line where the phases of the incident and reflected coincide, the voltage is maximum and equal to Ul max -= Ulo(1 + r) = Ulo(1 + 0,2) = 1,2Ulo, and in those where the phases are opposite, it is minimal and equal to Ul min = Ulo (1 - 0,2) = = 0,8 Ulo. By definition, SWR \u1d Ul max / / Ul min \u2d 0l8Ulo / 1I5Ulo \uXNUMXd XNUMXIXNUMX.

The formulas for calculating SWR and r can also be written as: SWR = (1+r)/(1-r) and r = = (SWR-1)/(SWR+1). We note an important point - the sum of the maximum and minimum voltages Ul max + Ul min = Ulo (1 + r) + Ulo (1 - r) = 2Uno, and their difference Ul max - Ul min = 2Ulo. Based on the obtained values, it is possible to calculate the power of the incident wave Рpad = Ulo2/Zo and the power of the reflected wave Pref = (rUlo)2/Zo. In our case (for SWR = 1,5 and r = 0,2), the power of the reflected wave will be only 4% of the power of the incident wave.

Determining the SWR by measuring the voltage distribution along the line section in search of the values ​​of Ul max and Ul min was widely used in the past

not only on open overhead lines, but also in coaxial feeders (mainly on VHF). To do this, we used the measuring section of the feeder, which has a long longitudinal slot, along which a trolley moved with a probe inserted into it - the head of an RF voltmeter.

SWR can be determined by measuring the current Il in one of the wires of the line in a section less than 0,5λ long. Having determined the maximum and minimum values, calculate SWR \uXNUMXd Imax / Imin. To measure the current, a current-to-voltage converter is used in the form of a current transformer (TT) with a load resistor, the voltage across which is proportional and in-phase to the measured current. We note an interesting fact - with certain parameters of the TT, at its output it is possible to obtain a voltage equal to the voltage on the line (between the conductors), i.e. Utl = IlZo.

On fig. 1b shows together a graph of the change in Ul along the line and a graph of the change in Utl. Graphs have the same amplitude and shape, but are shifted relative to each other by 0.25X. An analysis of these curves shows that it is possible to determine g (or SWR) by simultaneously measuring the values ​​of Ul and UTL at any point on the line. At the locations of the maxima and minima of both curves (points 1 and 2), this is obvious: the ratio of these values ​​Ul / Utl (or Utl / Ul) is equal to SWR, the sum is 2Ulo, and the difference is 2rUlo. At intermediate points, Ul and Utl are phase-shifted, and they must be added already as vectors, however, the above relationships are preserved, since the reflected voltage wave is always inverse in phase to the reflected current wave, and rUlo = rUtlo.

Therefore, a device containing a voltmeter, a calibrated current-to-voltage converter and an addition-subtraction circuit will make it possible to determine such line parameters as r or SWR, as well as Ppad and Rotr when it is turned on anywhere in the line.

The first information about devices of this kind dates back to 1943 and is reproduced in [1]. The first practical devices known to the author were described in [2, 3]. The variant of the circuit taken as a basis is shown in Fig. 2. The device contained:

• voltage sensor - a capacitive divider into C1 and C2 with an output voltage Uc, much less than the voltage on the line Ul. The ratio p \uXNUMXd Uc / Ul is called the coupling coefficient;
• current transformer T1, wound on a carbonyl ring magnetic circuit. Its primary winding had one turn in the form of a conductor passing through the center of the ring, the secondary - n turns, the load on the secondary winding - resistor R1, the output voltage - 2Ut. The secondary winding can be made of two separate windings with voltage Ut each and with its own load resistor, however, it is structurally more convenient to make one winding with a tap from the middle;
• detectors on diodes VD1 and VD2, switch SA1 and voltmeter on microammeter RA1 with additional resistors.

The secondary winding of the transformer T1 is connected in such a way that when the transmitter is connected to the left connector according to the circuit, and the load is connected to the right one, the total voltage Uc + UT is supplied to the diode VD1, and the difference voltage is applied to the diode VD2. When a resistive reference load with a resistance equal to the wave impedance of the line is connected to the output of the SWR meter, there is no reflected wave and, therefore, the RF voltage on VD2 can be zero. This is achieved in the process of balancing the device by equalizing the voltages UT and Uc using a tuning capacitor C1. As shown above, after such a setting, the magnitude of the difference voltage (at Zн≠Zо) will be proportional to the reflection coefficient r. Measurements with a real load are performed as follows. First, in the position of the SA1 switch shown in the diagram ("Incident wave"), the calibration variable resistor R3 sets the instrument needle to the last division of the scale (for example, 100 μA). Then the switch SA1 is moved to the lower position according to the scheme ("Reflected wave") and the value of r is counted. For the case with RH = 75 Ohm, the device should show 20 μA, which corresponds to r = 0,2. The SWR value is determined by the above formula - SWR \u1d (0,2 +1) / / (0,2-1,5) \u100d 20 or SWR \u100d (20 + 1,5) / / (XNUMX-XNUMX) \uXNUMXd XNUMX. In this example, the detector is assumed to be linear - in fact, it is necessary to introduce a correction that takes into account its non-linearity. When properly calibrated, the instrument can be used to measure incident and reflected powers.

The accuracy of the SWR meter as a measuring device depends on a number of factors, primarily on the accuracy of the device balancing in the SA1 "Reflected wave" position at Rн = Zo. Ideal balancing corresponds to voltages Uс and Uт, equal in magnitude and strictly opposite in phase, i.e. their difference (algebraic sum) is equal to zero. In a real design, there is always an unbalanced balance Ures. Let's look at an example of how this affects the final measurement result. Let's assume that when balancing, the voltages Uc = 0,5 V and Ut = 0,45 V were obtained (i.e., the unbalance is 0,05 V, which is quite real). With a load of Rn = 75 Ohm in a 50-ohm line, we actually have SWR = 75/50 = 1,5 and r = 0,2, and the magnitude of the reflected wave, recalculated to the in-device levels, will be rUc = 0,2x0,5 = 0,1, 0,2 V and rUt = 0,45x0,09 = XNUMX V.

Let's turn again to Fig. 1b, the curves on which are given for SWR = 1,5 (the curves Ul and Utl for the line will in our case correspond to Uc and Ut). At point 1 Uс max = 0,5 + 0,1 = 0,6 V, Ut min = 0,45 - 0,09 = 0,36 V and SWR = 0,6 / 0,36 = 1,67. At the point 2UTmax = 0,45 + 0,09 = 0,54 V, Ucmin = 0,5 - 0,1 = 0.4 and SWR = 0,54 / 0,4 = 1,35. From this simple calculation, it can be seen that, depending on the place where such an SWR meter is connected to a line with a real SWR = 1,5, or when the line length between the device and the load changes, different SWR values ​​can be read - from 1,35 to 1,67!

What can lead to inaccurate balancing?

1. The presence of a cutoff voltage of the germanium diode (in our case, VD2), at which it ceases to conduct, is approximately 0,05 V. Therefore, with UOCT < 0,05 V, the PA1 device will show "zero" and a balancing error can be made. The relative inaccuracy will decrease significantly if the voltages Uc and, accordingly, UT are raised several times. For example, with Uc = 2 V and UT = 1,95 V (Ures = 0,05 V), the SWR limits for the above example will only be from 1,46 to 1,54.

2. Presence of frequency dependence of voltages Uc or UT. In this case, precise balancing can not be achieved in the entire operating frequency range. Let's look at an example of one of the possible reasons. Suppose the device uses a divider capacitor C2 with a capacity of 150 pF with wire leads 0,5 mm in diameter and 10 mm long each. The measured inductance of a wire of this diameter 20 mm long turned out to be L = 0,03 μH. At the upper operating frequency f = 30 MHz, the capacitor resistance will be Xc = 1 / 2πfС = -j35,4 Ohm, the total reactance of the terminals XL = 22πfL = j5,7 Ohm. As a result, the resistance of the lower arm of the divider will decrease to -j35,4 + j5f7 = -j29,7 Ohm (it corresponds to a 177 pF capacitor). At the same time, at frequencies from 7 MHz and below, the influence of the leads is negligible. Hence the conclusion - in the lower arm of the divider, non-inductive capacitors with minimal leads (for example, reference or feed-through) should be used and several capacitors should be connected in parallel. The conclusions of the "upper" capacitor C1 practically do not affect the situation, since the Xc of the upper capacitor is several tens of times greater than that of the lower one. It is possible to obtain uniform balancing in the entire operating frequency band using an original solution, which will be discussed in the description of practical designs.

3. The influence of parasitic reactivity leads to the out-of-phase voltages Uc and UT (at ZH = Zo!). A phase shift of several degrees slightly affects their sum, but greatly worsens the balance. For example, if the phase shift is only α = 3° and Uc = UT = 2 V, the unbalanced balance will be Ures - Ucsinα = 2x0,052 = 0,104 V. Let's consider the possible reasons for this effect.

3.1. Influence of the reactivity of the outputs of the secondary winding. With a lead length of only 10 mm at the upper limit of the KB range, their resistance XL = j5,7 Ohm (see the previous example) and the phase of the current in the secondary circuit T1 will be shifted by an angle α = in relation to the current in the line (and voltage Uc) arctan(XL/R1). Here R1 is the load resistance of the transformer, which usually ranges from 10 to 100 ohms. For extreme values, we get α = arctan(5,7/10) = 30° (!) and α = arctan(5,7/100) - 3°. In fact, the parasitic inductance in the secondary circuit can be even greater due to the presence of leakage inductance T1 and lead inductance R1. Note that although the impedance of the secondary circuit increases at higher frequencies, the voltage UT taken directly from R1 remains unchanged (see below for the property of the current transformer).

3.2. The inductive resistance of the secondary winding T1 at the lower frequencies of the operating range (~ 1,8 MHz) can significantly shunt R1, which will lead to a decrease in UT and its phase shift.

3.3. Resistance R2 is part of the detector circuit. Since, according to the scheme, it shunts C2, at lower frequencies the division factor can receive frequency and phase dependences.

3.4. In the scheme of Fig. 2 detectors on VD1 or VD2 in the open state shunt the lower arm of the capacitive divider on C2 with their input resistance RBX, i.e. RBX acts in the same way as R2. The influence of RBX is insignificant at (R3 + R2) more than 40 kOhm, which requires the use of a sensitive indicator RA1 with a total deviation current of not more than 100 μA and an RF voltage on VD1 of at least 4 V.

3.5. The input and output connectors of the SWR meter are usually spaced 30...100 mm apart. At a frequency of 30 MHz, the voltage phase difference at the connectors will be α= [(0,03... 0,1)/10]360°- 1... 3,5°. How this can affect performance is shown in Fig. 3a and fig. 3b. The difference between the circuits in these figures is only that the capacitor C1 is connected to different connectors (T1 in both cases is in the middle of the conductor between the connectors).

In the first case, the uncompensated residual can be reduced if the UOCT phase is corrected using a small parallel-connected capacitor Sk, and in the second case, by connecting a small inductance Lk in the form of a wire loop in series with R1. This method is often used in both home-made and "proprietary" SWR meters, but this should not be done. To verify this, it is enough to turn the device so that the input connector becomes the output. At the same time, the compensation that helped before the turn will become harmful - Uoct will increase significantly. When working on a real line with an inconsistent load, depending on the length of the line, the device can get to a place on the line where the correction introduced will "improve" the real SWR or, conversely, "worse" it. Either way, it will be incorrect. The recommendation is to place the connectors as close as possible to each other and use the original circuit design below.

To illustrate how strongly the reasons discussed above can affect the reliability of the SWR meter readings, in Fig. Figure 4 shows the results of checking two factory-made devices [4]. The check consisted in the fact that an unmatched load with a calculated SWR = 2,25 was installed at the end of the line, consisting of a number of serially connected cable segments with Zо = 50 Ohm, each λ/8 long.

During measurements, the total length of the line varied from λ/8 to 5/8λ. Two devices were tested: inexpensive BRAND X (curve 2) and one of the best models - BIRD 43 (curve 3). Curve 1 shows true SWR. As they say, comments are superfluous.

On fig. Figure 5 shows a graph of the dependence of the measurement error on the magnitude of the directivity D (directivity) of the SWR meter [4]. Similar plots for KBV = 1/SWR are given in [5]. With regard to the design of Fig. 2, this coefficient is equal to the ratio of the RF voltages on the diodes VD1 and VD2 when connected to the output of the load SWR meter Rn = Zo D = 20lg (2Uo / Ures). Thus, the better it was possible to balance the circuit (the smaller Ures), the higher D. You can also use the readings of the indicator PA1 - D = 20 x x lg (Ifall / Iotp). however, this D value will be less accurate due to the non-linearity of the diodes.

On the graph, the real SWR values ​​are plotted along the horizontal axis, and the measured ones, taking into account the error, depending on the D value of the SWR meter, are plotted on the vertical axis. The dotted line shows an example - real SWR \u2d 20, a device with D \u1,5d 2,5 dB will give readings of 40 or 1,9, and with D \u2,1d XNUMX dB - XNUMX or XNUMX, respectively.

As follows from the literature data [2, 3], the SWR meter according to the scheme of Fig. 2 has D - 20 dB. This means that without significant correction it cannot be used for accurate measurements.

The second most important reason for incorrect SWR readings is related to the nonlinearity of the current-voltage characteristic of the detector diodes. This leads to a dependence of the readings on the level of supplied power, especially in the initial part of the PA1 indicator scale. In branded SWR meters, two scales are often made on the indicator - for low and high power levels.

The current transformer T1 is an important part of the SWR meter. Its main characteristics are the same as those of a more familiar voltage transformer: the number of turns of the primary winding n1 and secondary n2, the transformation ratio k \u2d n1 / n2, the secondary winding current I1 \u1d l2 / k. The difference is that the current through the primary winding is determined by the external circuit (in our case, this is the current in the feeder) and does not depend on the load resistance of the secondary winding R1, so the current l50 also does not depend on the resistance value of the resistor R100. For example, if power P = 1 W is transmitted through the feeder Zo = XNUMX Ohm, current IXNUMX = √P/Zo\u1,41d 20 A and at k \u2d 1, the current of the secondary winding will be l0,07 \u1d I2 / k - 2 A. The voltage at the terminals of the secondary winding will be determined by the value of R1: 1UT \u68d l2 x R4,8 and at R2 \u2d 1 Ohms it will be 0,34UT \u1d 1 V. The power dissipated on the resistor P \uXNUMXd (XNUMXUT) XNUMX / RXNUMX \uXNUMXd XNUMX W. Let's pay attention to the peculiarity of the current transformer - the fewer turns in the secondary winding, the greater the voltage at its terminals (for the same RXNUMX). The most difficult mode for a current transformer is the idle mode (RXNUMX = ∞), while the voltage at its output increases sharply, the magnetic circuit is saturated and heated up so much that it can collapse.

In most cases, one turn is used in the primary winding. This coil can have different shapes, as shown in Fig. 6a and fig. 6, b (they are equivalent), but the winding according to fig. 6, in - this is already two turns.

A separate issue is the use of a screen connected to the body in the form of a tube between the central wire and the secondary winding. On the one hand, the screen eliminates the capacitive coupling between the windings, which somewhat improves the balance of the difference signal; on the other hand, eddy currents appear in the screen, which also affect the balancing. Practice has shown that with and without a screen, you can get approximately the same results. If the screen is still used, its length should be made minimal, approximately equal to the width of the applied magnetic circuit, and connected to the body with a wide short conductor. "Grounding" the screen should be done on the middle line, equidistant from both connectors. For the screen, you can use a brass tube with a diameter of 4 mm from telescopic antennas.

For SWR meters for a through power of up to 1 kW, ferrite ring magnetic circuits with dimensions K12x6x4 and even K10x6x3 are suitable. Practice has shown that the optimal number of turns is n2 = 20. With a secondary winding inductance of 40 ... 60 μH, the greatest frequency uniformity is obtained (the permissible value is up to 200 μH). It is possible to use magnetic circuits with a permeability from 200 to 1000, while it is desirable to choose a size that will provide the optimal winding inductance.

It is possible to use magnetic circuits with lower permeability, if you apply larger sizes, increase the number of turns and / or reduce the resistance R1. If the permeability of the existing magnetic circuits is unknown, it can be determined with an inductance meter. To do this, wind ten turns on an unknown magnetic circuit (each intersection of the core inner hole with a wire is considered a turn), measure the coil inductance L (μH) and substitute this value into the formula μ = 2,5 LDavg/S, where Dav is the average diameter of the magnetic circuit in cm ; S - core section in cm² (example - for K10x6x3 Dcp = 0,8 cm and S = 0,2x0,3 = 0,06 cm²).

If μ of the magnetic circuit is known, the inductance of the winding of n turns can be calculated: L = μn²S/250Dcp.

The applicability of magnetic circuits to a power level of 1 kW or more can be checked even at 100 W in the feeder. To do this, temporarily install a resistor R1, 4 times larger, respectively, the voltage Ut will also increase 4 times, and this is equivalent to an increase in the transmitted power by 16 times. The heating of the magnetic circuit can be checked by touch (the power on the temporary resistor R1 will also increase by 4 times). In real conditions, the power on the resistor R1 increases in proportion to the growth of power in the feeder.

SWR meters UT1MA

The two designs of the UT1MA SWR meter, which will be discussed below, have almost the same circuit, but different designs. In the first version (KMA - 01), the high-frequency sensor and the indicator part are separate. The sensor has input and output coaxial connectors and can be installed anywhere in the feeder path. It is connected to the indicator with a three-wire cable of any length. In the second variant (KMA - 02) both units are located in one housing.

The diagram of the SWR - meter is shown in fig. 7 and differs from the basic circuit in Fig. 2 by the presence of three correction circuits.

Let's consider these differences.

1. The upper arm of the capacitive divider C1 is made of two identical constant capacitors C1 = C1' + C1 ", connected respectively to the input and output connectors. approaches the UT phase, which improves the balance of the instrument.
2. Due to the introduction of the L1 coil, the resistance of the upper arm of the capacitive divider becomes frequency-dependent, which makes it possible to equalize the balancing at the upper edge of the operating range (21 ... 30 MHz).
3. By choosing the resistor R2 (i.e., the time constant of the R2C2 chain), you can compensate for the imbalance caused by the voltage drop UT and its phase shift at the lower end of the range (1,8 ... 3,5 MHz).

In addition, balancing is carried out by a trimmer capacitor included in the lower arm of the divider. This simplifies installation and allows the use of a low-power small-sized trimmer capacitor.

The design provides for the possibility of measuring the power of the incident and reflected waves. To do this, switch SA2 into the indicator circuit instead of a variable calibration resistor R4, a tuning resistor R5 is introduced, which sets the desired limit of the measured power.

The use of optimal correction and the rational design of the device made it possible to obtain a directivity factor D in the range of 35 ... 45 dB in the frequency band of 1,8 ... 30 MHz.

In SWR - meters, the following details are used.

The secondary winding of the transformer T1 contains 2 x 10 turns (winding in 2 wires) with a 0,35 PEV wire, placed evenly on a K12 x 6 x 4 ferrite ring with a permeability of about 400 (measured inductance ~ 90 μH).

Resistor R1 - 68 ohm MLT, preferably without a helical groove on the body of the resistor. With a passing power of less than 250 W, it is enough to install a resistor with a dissipation power of 1 W, with a power of 500 W - 2 W. With a power of 1 kW, the resistor R1 can be made up of two resistors connected in parallel with a resistance of 130 ohms and a power of 2 W each. However, if the COP V-meter is designed for a high power level, it makes sense to double the number of turns of the secondary winding T1 (up to 2 x 20 turns). This will reduce the required power dissipation of the resistor R4 by 1 times (in this case, the capacitor C2 should have twice the capacitance).

The capacitance of each of the capacitors C G and C1 "can be within 2,4 ... 3 pF (KT, KTK, KD for an operating voltage of 500 V at P ≥ 1 kW and 200 ... 250 V at lower power). Capacitors C2 - for any voltage (KTK or other non-inductive, one or 2 - 3 in parallel), capacitor C3 - small-sized trimmer with capacitance change limits of 3 ... 20 pF (KPK - M, KT - 4). The required capacitance of capacitor C2 depends on the total value of the capacitance of the upper arm of the capacitive divider, which includes, in addition to the capacitors C' + C1 ", also the capacitance C0 ~ 1 pF between the secondary winding of the transformer T1 and the central conductor. The total capacitance of the lower arm - C2 plus C3 at R1 = 68 ohms should be about 30 times the capacitance of the upper one. Diodes VD1 and VD2 - D311, capacitors C4, C5 and C6 - with a capacity of 0,0033 ... 0,01 μF (KM or other high-frequency), indicator RA1 - M2003 with a total deviation current of 100 μA, variable resistor R4 - 150 kOhm SP - 4 - 2m, trimmer resistor R4 - 150 kOhm. Resistor R3 has a resistance of 10 kOhm - it protects the indicator from possible overload.

The value of the corrective inductance L1 can be determined as follows. When balancing the device (without L1), it is necessary to note the positions of the rotor of the tuning capacitor C3 at frequencies of 14 and 29 MHz, then unsolder it and measure the capacitance in both marked positions. Let's say for the upper frequency the capacitance turned out to be less by 5 pF, and the total capacitance of the lower arm of the divider is about 130 pF, i.e. the difference is 5/130 or about 4%. Therefore, for frequency equalization, it is necessary to reduce the resistance of the upper arm at a frequency of 29 MHz by ~ 4% as well. For example, at C1 + C0 = 5 pF capacitance Xc = 1/2πfС - j1100 Ohm, respectively, Xc - j44 Ohm and L1 = XL1 / 2πf = = 0,24 μH.

In the author's devices, the L1 coil had 8 ... 9 turns with a PELSHO 0,29 wire. The inner diameter of the coil is 5 mm, the winding is dense, followed by impregnation with BF-2 glue. The final number of turns is specified after it is installed in place. Initially, balancing is performed at a frequency of 14 MHz, then the frequency is set to 29 MHz and the number of turns of the coil L1 is selected, at which the circuit is balanced at both frequencies at the same position of the trimmer C3.

After achieving good balancing at medium and high frequencies, a frequency of 1,8 MHz is set, a variable resistor with a resistance of 2 ... 15 kOhm is temporarily soldered in place of the resistor R20 and a value is found at which UOCT is minimal. The resistance value of the resistor R2 depends on the inductance of the secondary winding T1 and lies within 5 ... 20 kOhm for its inductance of 40 ... 200 μH (higher resistance values ​​\uXNUMXb\uXNUMXbfor greater inductance).

In amateur radio conditions, most often a microammeter with a linear scale is used in the SWR meter indicator and the reading is carried out according to the formula SWR \u7d (Ipad + Iotr) / (Ipad -Iotr), where I in microamperes is the indicator reading in the "falling" and "reflected" modes respectively. This does not take into account the error due to the nonlinearity of the initial section of the CVC of the diodes. A test using loads of various sizes at a frequency of 100 MHz showed that at a power of about 1 W, the indicator readings were on average one division (25 μA) less than the real values, at 2,5 W - less by 3 ... 10 μA, and at 4 W - by 100 μA. Hence a simple recommendation: for the 10-watt version, shift the initial (zero) position of the instrument arrow one division up in advance, and when using 4 W (for example, when tuning an antenna), add another 100 μA to the reading on the scale in the “reflected” position. An example is the incident/reflected readings, respectively, 16/100 µA, and the correct SWR would be (20 + 100) / (20 - 1,5) = 500. With a significant power - XNUMX W or more - this correction is not necessary.

It should be noted that all types of amateur SWR meters (on a current transformer, bridge, on directional couplers) give values ​​for the reflection coefficient r, and the SWR value then has to be calculated. Meanwhile, it is r that is the main indicator of the degree of agreement, and SWR is a derivative indicator. This can be confirmed by the fact that in telecommunications the degree of agreement is characterized by attenuation of inconsistency (the same r, only in decibels). Expensive branded devices also provide a countdown r called return loss (return loss).

This remark is made in order to emphasize the following fact. In amateur conditions, it is quite difficult to make an indicator scale in SWR values, but r can be read directly on a linear scale.

What happens if silicon diodes are used as detectors? If a germanium diode at room temperature has a cutoff voltage at which the current through the diode is only 0,2 ... 0,3 μA, is about 0,045 V, then the silicon diode has already 0,3 V. silicon diodes, it is necessary to increase the voltage levels Uc and UT (!) by more than 6 times. In the experiment, when replacing diodes D311 with KD522 at P = 100 W, load Zn = 75 Ohm and the same Uc and UT, the following figures were obtained: before replacement - 100/19 and SWR = 1,48, after replacement - 100/12 and calculated SWR = 1,27. The use of a doubling circuit on KD522 diodes gave an even worse result - 100/11 and a calculated SWR = 1,25.

The sensor housing in a separate version can be made of copper, aluminum or soldered from plates of double-sided foiled fiberglass with a thickness of 1,5...2 mm. A sketch of such a design is shown in Fig. 8, a.

The case consists of two compartments, in one opposite each other there are HF connectors (CP - 50 or SO - 239 with flanges measuring 25x25 mm), a jumper made of a wire with a diameter of 1,4 mm in polyethylene insulation with a diameter of 4,8 mm (from the cable PK50 - 4), current transformer T1, capacitors of the capacitive divider and compensation coil L1, in the other - resistors R1, R2, diodes, tuning and blocking capacitors and a small-sized low-frequency connector. Conclusions T1 of the minimum length. The connection point of the capacitors C1 'and C1 "with the coil L1 "hangs in the air", and the connection point of the capacitors C4 and C5 of the middle output of the XZ connector is connected to the device case.

Partitions 2, 3 and 5 have the same dimensions. There are no holes in the partition 2, and in the partition 5 a hole is made for a specific low-frequency connector through which the indicator unit will be connected. In the middle jumper 3 (Fig. 8, b), foil is selected around three holes on both sides, and three through conductors are installed in the holes (for example, M2 and M1 brass screws). Sketches of sidewalls 4 and 8 are shown in fig. XNUMX, c. The dotted lines show the joints before soldering, which is made on both sides for greater strength and electrical contact.

The design of the indicator block without features is not considered here either.

The RF sensor of the second version of the SWR meter is mounted on a removable rear wall (copper, aluminum, brass) of the metal case of the SWR meter (Fig. 9).

Unlike the first option, all parts (except for T1 and connectors XW1 and XW2) are mounted on a printed circuit board (Fig. 10), a low-frequency connector of the type of interconnected television is soldered there.

Capacitors C1 'and C1 "are soldered to the contact pad on the printed circuit board at one end, and to the RF connectors at the other ends. Elements C2, C3 and L1 are located on the side of the foil. The limiting resistor R3 is transferred to the board (R3 'and R3" are shown in the diagram dotted line). Diodes VD1 and VD2 are installed vertically. The board is attached to the panel between the RF connectors using small soldered copper corners 0,5...1 mm thick (the soldering point is shown in Fig. 10 by a dotted line). It is desirable to cover the sensor with a screen. The design of the indicator - without features.

To set up and test the SWR meter, you need a 50 Ohm exemplary load resistor (antenna equivalent) with a power of 50 ... 100 W. One of the possible amateur radio designs is shown in Fig. 11. It uses a common TVO resistor with a resistance of 51 ohms and a dissipation power of 60 W (rectangle with dimensions of 45 x 25 x 180 mm).

Inside the ceramic body of the resistor is a long cylindrical channel filled with a resistive substance. The resistor should be firmly pressed against the bottom of the aluminum casing. This improves heat dissipation and creates a distributed capacitance that improves wide-bandwidth. With the help of additional resistors with a dissipation power of 2 W, the input load resistance is set within 49,9 ... 50,1 Ohm. With a small correction capacitor at the input (~ 10 pF), it is possible to obtain a load with an SWR of at least 1,05 in the frequency band up to 30 MHz based on this resistor. Excellent loads are obtained from special small-sized resistors of the P1 - 3 type with a nominal value of 49,9 ohms, which can withstand significant power when using an external radiator.

Comparative tests of SWR meters from different companies and devices described in this article were carried out. The test consisted in the fact that an unmatched load of 100 ohms (equivalent to a factory-made 50 W antenna) was connected through a tested 75-ohm SWR meter to a transmitter with an output power of about 100 W and two measurements were made. One - when connected with a short PK50 cable 10 cm long, the other - through a PK50 cable ~ 0,25λ long. The smaller the spread of readings, the more reliable the device.

At a frequency of 29 MHz, the following SWR values ​​were obtained:

• DRAKE WH - 7......1,46/1,54
• DIAMOND SX - 100......1,3/1,7
• ALAN KW-220......1,3/1,7
• ROGER RSM-600......1,35/1,65
• UT1MA......1,44/1,5

With a load of 50 ohms, for any length of cables, all devices "unanimously" showed SWR < 1,1.

The reason for the large spread in the readings of the RSM - 600 was found out during its study. In this device, not a capacitive divider is used as a voltage sensor, but a step-down voltage transformer with a fixed transformation ratio. This eliminates the "problems" of the capacitive divider, but reduces the reliability of the device when measuring high powers (the maximum power of the RSM - 600 is only 200/400 W). There is no tuning element in his circuit, so the load resistor of the current transformer must be of high accuracy (at least 50 ± 0,5 Ohm), but a resistor with a resistance of 47,4 Ohm was actually used. After replacing it with a 49,9 ohm resistor, the measurement results became much better - 1,48 / 1,58. Perhaps the same reason is associated with a large spread in the readings of the SX - 100 and KW - 220 instruments.

Measurement at an unmatched load with an optional 50 ohm quarter-wave cable is a reliable way to check the quality of an SWR meter. We note three points:

1. For such a test, you can also use a load of 50 ohms, if you connect a capacitor parallel to its input, for example, in the form of a small segment of a coaxial cable open at the end. Connection is conveniently made through a coaxial tee transition. Experimental data - with a segment of PK50 28 cm long at a frequency of 29 MHz, such a combined load had an SWR - 1,3, and with a length of 79 cm - SWR - - 2,5 (connect any load to the SWR - meter only with a 50 - ohm cable) .
2. The real SWR in the line approximately corresponds to the average of two read values ​​(with and without an additional quarter-wave cable).
3. When measuring a real antenna-feeder device, difficulties may arise due to the current flowing to the outer surface of the cable sheath. In the presence of such a current, changing the length of the feeder from below can lead to a change in this current, which will lead to a change in the load of the feeder and the actual SWR. It is possible to reduce the influence of external current by turning the feeder entering the room in the form of a bay of 15 ... 20 turns with a diameter of 15 ... 20 cm (protective choke).

Literature

1. D. Lechner, P. Finck. Kurzwellen sender. - Berlin: Militarverlag, 1979.
2. WB Bruene- An Inside Pictures of Directional Wattmeters. - QST, April, 1959.
3. D. DeMaw. In-Line RF Power Metering. - QST, December, 1969.
4. W. Orr, S. Cowan. The beam antenna handbook. - RAC, USA, 1993.
5. Beketov V., Kharchenko K. Measurements and tests in the design and adjustment of amateur radio antennas. - M.: Communication, 1971.

Author: E.Gutkin (UT1MA), Lugansk, Ukraine

See other articles Section Antennas. Measurements, adjustment, coordination.

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