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ENCYCLOPEDIA OF RADIO ELECTRONICS AND ELECTRICAL ENGINEERING Improvement of the capacitance and inductance meter. Encyclopedia of radio electronics and electrical engineering
Encyclopedia of radio electronics and electrical engineering / Measuring technology Simple capacitance and inductance meters, like those described in [1, 2], have a low measurement accuracy. To understand its causes, consider the principle of measurement, which explains Fig. 1. When measuring the capacitance (Fig. 1, a), the capacitor Cx from the voltage source U receives a charge q \u1d U·CX, and after switching by means of the switch S, the discharge current flows through the measuring device. Measurement of the inductance (Fig. 1b) is also based on the registration of the discharge current flowing through the measuring circuit. If we accept the switching as instantaneous, then the charge is determined here by the ratio of the magnetic flux in the inductance, equal to I Lx, to the total resistance of the DC circuit R and + RL, i.e. q \uXNUMXd XNUMX-Lx / (R and + RL) In practice, switching is carried out periodically with a frequency f using electronic switches, and the measuring device registers the direct component of the current Ii = q -f.
The first reason for measurement errors in the described devices is related to the insufficient sensitivity of the microammeter that measures the current Ii. Because of this, the switching frequency f has to be chosen high, and the capacitor Cx after it is disconnected from the measuring circuit still retains a significant part of the initial charge q, which somewhat reduces the actually measured current Ii. This decrease depends on the capacitance of the capacitor: the smaller it is, the more complete the discharge of the capacitor. Therefore, the scale of the measuring instrument must be non-linear, and using the microammeter's own linear scale can lead to an error of several percent. In the case of inductance measurements, in addition to the error due to the high switching frequency and the associated non-linearity, an additional error occurs for coils with a noticeable winding resistance RL. If, for example, the device is calibrated against a reference inductance with its own resistance RL, much less than Ri, and then the coil inductance is measured with a resistance RL commensurate with R, then the readings will be underestimated by (R and + RL) / R and times. It is sometimes necessary to take into account active resistance when calibrating using reference chokes, since, for example, a DM-0,1 choke with an inductance of 500 μH has RL = 10 Ohm. To eliminate the noted sources of error, the measuring part of the device from [2] was changed (Fig. 2). Thanks to the use of op amp DA1, the sensitivity of the meter is increased by 10 times in terms of current, and the switching frequency is reduced by the same amount at the corresponding limits. As a result, the non-linearity of the scale became less than 1%.
The upper limits of measuring capacitance and inductance at a switching frequency of 1 MHz with an M24 microammeter at 100 μA are 10 pF and 1 μH, respectively. The reduction in mounting capacity is achieved by introducing an additional third clamp for measured coils and capacitors and eliminating the L-C switch. In addition, the switching diodes VD1-VD3 are soldered by one of the leads directly to the terminals. As a result, with free clamps, the mounting capacitance, which can be judged by the deviation of the arrow from zero, is less than 1 pF. The switching frequency within 10 uF and 1 H is very low and amounts to 1 Hz. In this case, the inertia of the microammeter is insufficient to smooth out the fluctuations of the arrow, and therefore the capacitance of the capacitor C2 is chosen to be 4700 μF. When measuring at this frequency, the pointer settling time increases to tens of seconds. At other limits with a higher switching frequency, a capacitance of about 470 μF is sufficient, and then the measurement time is seconds. On the switch of measurement limits, it is advisable to add a contact group that includes the full capacity of C2 only at this last limit. u= R1 + R2. With a significant resistance of the winding, the value of the introduced (right) part of R1 should be reduced so that the total value R and = RL + R1 + R2 remains unchanged. If a precision resistor is available, it may be provided with a graduated scale. The design uses a conventional resistor SP2-3b, and therefore sockets XS4, XS5 are added to measure the output part of R1 with an ohmmeter used to measure the resistance of the winding. To switch the elements under test, a complementary emitter follower on transistors VT1, VT2 is used to the power source, to the bases of which voltage pulses in the form of a meander are fed through the parallel-connected elements R5, C5. The required switching frequency is set by a quartz resonator oscillator and a sequence of decimal divider counters made on K176 or K561 series microcircuits. This part of the scheme did not differ in any way from that given in [2] and is therefore omitted here. So that fluctuations in the supply voltage do not introduce an additional error into the measurements, a voltage of +9 V is supplied to this part of the circuit and to the switch from the stabilizer. The power supply of the op amp DA1 is allowed from a power source with unstabilized voltages of ±12 V; to eliminate interference from the pulse shaper, capacitors C3, C4 are added to the power circuit, placed near this microcircuit. Setting up the meter comes down to zeroing the measuring device with resistor R4 at one of the largest limits ("1 μF" or "0,1 μF"), calibrating by reference capacitor with adjustment by resistor R3, and then by reference inductance with adjustment by R2 (at this engine of the resistor R1 sets its resistance between XS4 and XS5, equal to the resistance of the coil winding). Trimmer resistors R2, R3 are preferably multi-turn (SP5-2, SP5-22, etc.). Literature
Author: V.Ivanov, Rostov-on-Don
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